Load modulation amplifier

ABSTRACT

A load modulation amplifier is disclosed. The load modulation amplifier includes a carrier amplifier for amplifying a radio frequency signal when input power of the radio frequency signal is below a predetermined power threshold value and a peak amplifier coupled in parallel with the carrier amplifier for amplifying the radio frequency signal when input power of the radio signal is above the predetermined power threshold value. The load modulation amplifier further includes an output quadrature coupler configured to combine power from both the carrier amplifier and the peak amplifier for output through an output load terminal. Output impedance of the peak amplifier monotonically increases with increasing output power at the output load terminal.

RELATED APPLICATIONS

This application is related to U.S. Pat. No. 9,484,865, issued Nov. 1,2016, titled RECONFIGURABLE LOAD MODULATION AMPLIFIER; U.S. patentapplication Ser. No. 15/278,450, filed Sep. 28, 2016, titledRECONFIGURABLE LOAD MODULATION AMPLIFIER; and U.S. patent applicationSer. No. 15/278,270, filed Sep. 28, 2016, titled RECONFIGURABLE LOADMODULATION AMPLIFIER, the disclosures of which are hereby incorporatedherein by reference in their entireties.

FIELD OF THE DISCLOSURE

The present disclosure pertains to amplifiers and in particular to loadmodulation amplifiers having a carrier amplifier and a peak amplifiercoupled in parallel.

BACKGROUND

Traditional Doherty power amplifiers have been employed to improve highpower backed off efficiency over a wide power range. However, asfrequency increases, Doherty power amplifier performance degradesoverall due to amplifier device parasitic capacitance and inductance. Ataround 15 GHz, backed off efficiency of Doherty power amplifiers beginsto decay linearly with frequency. Therefore, challenges remain withregard to maintaining desirable output backed off efficiency atfifth-generation (5G) wireless network millimeter wave frequencies thatinclude 28 GHz, 38 GHz, and 60 GHz. Moreover, 5G wireless networksemploying phased array applications are constrained by cost, complexity,and output power linearity. Further still, digital pre-distortiontechniques are undesirable as solutions to non-linear Doherty operationat millimeter wave frequencies. Therefore, a need remains for a loadmodulation power amplifier that provides both output power backed offefficiency and linear operation without digital pre-distortion foroperation at 5G wireless network millimeter wave frequencies.

SUMMARY

A load modulation amplifier is disclosed. The load modulation amplifierincludes a carrier amplifier for amplifying a radio frequency signalwhen input power of the radio frequency signal is below a predeterminedpower threshold value and a peak amplifier coupled in parallel with thecarrier amplifier for amplifying the radio frequency signal when inputpower of the radio signal is above the predetermined power thresholdvalue. The load modulation amplifier further includes an outputquadrature coupler configured to combine power from both the carrieramplifier and the peak amplifier for output through an output loadterminal. Output impedance of the peak amplifier monotonically increaseswith increasing output power at the output load terminal.

Those skilled in the art will appreciate the scope of the presentdisclosure and realize additional aspects thereof after reading thefollowing detailed description of the preferred embodiments inassociation with the accompanying drawing figures.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

The accompanying drawing figures incorporated in and forming a part ofthis specification illustrate several aspects of the disclosure and,together with the description, serve to explain the principles of thedisclosure.

FIG. 1 is a schematic of a first embodiment of a load modulationamplifier that is structured and configured in accordance with thepresent disclosure.

FIG. 2 is a schematic of a second embodiment of the load modulationamplifier that is structured and configured in accordance with thepresent disclosure.

FIG. 3 is an exemplary circuit topology for the matching network.

FIG. 4 is a graph of output impedance of the peak amplifier versusoutput power of the load modulation amplifier embodiments of the presentdisclosure.

FIG. 5 is a graph of carrier amplifier to output coupling versusmillimeter wave frequencies for the load modulation amplifierembodiments of the present disclosure.

FIG. 6 is a graph of carrier amplifier to output phase shift versusmillimeter wave frequencies for the load modulation amplifierembodiments of the present disclosure.

FIG. 7 is a graph of peak amplifier to output coupling versus millimeterwave frequencies for the load modulation amplifier embodiments of thepresent disclosure.

FIG. 8 is a graph of peak amplifier to output phase shift versusmillimeter wave frequencies for the load modulation amplifierembodiments of the present disclosure.

FIG. 9 is a graph of power added efficiency and drain efficiency versusoutput power for the second embodiment of the load modulation amplifierof FIG. 2.

FIG. 10 is a graph of third-order intermodulation distortion (IM3) andlinearity figure of merit (LFOM) as a function of output power for thesecond embodiment of the load modulation amplifier of FIG. 2.

FIG. 11 is a graph of gain delta versus output power for amplitudemodulation-amplitude modulation (AM-AM) distortion for the secondembodiment of the load modulation amplifier of FIG. 2 versus aconventional Doherty amplifier.

FIG. 12 is a graph of phase delta versus output power for AM-phasemodulation (AM-PM) distortion for the second embodiment of the loadmodulation amplifier of FIG. 2 versus a conventional Doherty amplifier.

FIG. 13 is a graph of error vector magnitude (EVM) versus output powerfor the second embodiment of the load modulation amplifier of FIG. 2 incomparison with a conventional Doherty amplifier.

FIG. 14 is a graph of EVM under a 2:1 voltage standing wave ratio (VSWR)mismatch versus output power for the second embodiment of the loadmodulation amplifier of FIG. 2.

FIG. 15 is a graph of EVM under a 2:1 VSWR mismatch versus output powerfor conventional Doherty amplifier.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the embodiments andillustrate the best mode of practicing the embodiments. Upon reading thefollowing description in light of the accompanying drawing figures,those skilled in the art will understand the concepts of the disclosureand will recognize applications of these concepts not particularlyaddressed herein. It should be understood that these concepts andapplications fall within the scope of the disclosure and theaccompanying claims.

It will be understood that, although the terms first, second, etc. maybe used herein to describe various elements, these elements should notbe limited by these terms. These terms are only used to distinguish oneelement from another. For example, a first element could be termed asecond element, and, similarly, a second element could be termed a firstelement, without departing from the scope of the present disclosure. Asused herein, the term “and/or” includes any and all combinations of oneor more of the associated listed items.

It will be understood that when an element such as a layer, region, orsubstrate is referred to as being “on” or extending “onto” anotherelement, it can be directly on or extend directly onto the other elementor intervening elements may also be present. In contrast, when anelement is referred to as being “directly on” or extending “directlyonto” another element, there are no intervening elements present.Likewise, it will be understood that when an element such as a layer,region, or substrate is referred to as being “over” or extending “over”another element, it can be directly over or extend directly over theother element or intervening elements may also be present. In contrast,when an element is referred to as being “directly over” or extending“directly over” another element, there are no intervening elementspresent. It will also be understood that when an element is referred toas being “connected” or “coupled” to another element, it can be directlyconnected or coupled to the other element or intervening elements may bepresent. In contrast, when an element is referred to as being “directlyconnected” or “directly coupled” to another element, there are nointervening elements present.

Relative terms such as “below” or “above” or “upper” or “lower” or“horizontal” or “vertical” may be used herein to describe a relationshipof one element, layer, or region to another element, layer, or region asillustrated in the Figures. It will be understood that these terms andthose discussed above are intended to encompass different orientationsof the device in addition to the orientation depicted in the Figures.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the disclosure.As used herein, the singular forms “a,” “an,” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises,”“comprising,” “includes,” and/or “including” when used herein specifythe presence of stated features, integers, steps, operations, elements,and/or components, but do not preclude the presence or addition of oneor more other features, integers, steps, operations, elements,components, and/or groups thereof.

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this disclosure belongs. It willbe further understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.

FIG. 1 is a schematic of a first embodiment of a load modulationamplifier 10 that is structured and configured in accordance with thepresent disclosure. In exemplary embodiments both an input quadraturecoupler 12 and an output quadrature coupler 14 each have four ports andare of the Lange type having microstrip or strip-line construction withgeometric symmetry that ensures quadrature power combining of the outputpower of a carrier amplifier 16 and a peak amplifier 18. The carrieramplifier 16 and the peak amplifier 18 are coupled in parallel by way ofthe input quadrature coupler 12 at an input terminal 20 labeled RF INand by way of the output quadrature coupler 14 at an output loadterminal 22 labeled RF OUT.

The input quadrature coupler 12 and the output quadrature coupler 14both typically have less than 0.25 dB of insertion loss and anapproximate octave frequency operating bandwidth. For example, in oneembodiment the input quadrature coupler 12 and the output quadraturecoupler 14 are both Lange couplers having a minimum frequency of 12 GHzand a maximum frequency of 24 GHz. In another embodiment, the inputquadrature coupler 12 and the output quadrature coupler 14 are bothLange couplers having a minimum frequency of 18 GHz and a maximumfrequency of 36 GHz. In yet another embodiment, the input quadraturecoupler 12 and the output quadrature coupler 14 are both Lange couplershaving a minimum frequency of 27 GHz and a maximum frequency of 54 GHz.

An input impedance termination network 24 is coupled between an inputtermination port of the input quadrature coupler 12 and ground. Anetwork output terminal 26 of a matching network 28 included with thepeak amplifier 18 is coupled to a first port P1 of the output quadraturecoupler 14. In at least one embodiment, the matching network 28 is madeup of only passive electrical components having inductance, capacitance,and resistance. An isolation termination network 30 having fixedimpedance that is higher than fixed impedance of the input impedancetermination network 24 is coupled between a second port P2 of the outputquadrature coupler 14 and ground. A bias current IBIAS for the peakamplifier 18 is set and/or controlled in combination with matchingimpedance of the matching network 28 such that an output impedance Z0 ofthe peak amplifier 18 seen at the network output terminal 26 increasesmonotonically with increasing output power of load modulation amplifier10. In this exemplary embodiment, the bias current IBIAS is supplied tothe peak amplifier 18 through the matching network 28.

In an exemplary embodiment, fixed impedance of the input impedancetermination network 24 is 50Ω and the fixed impedance of the isolationtermination network 30 is substantially greater than 50Ω, and in someexemplary embodiments, the fixed impedance of the isolation terminationnetwork 30 is on the order of 1000Ω. In the exemplary embodiment of FIG.1, an amplified radio frequency signal that is output from the carrieramplifier 16 is input into a third port P3 and undergoes a 0° phaseshift before exiting a fourth port P4 that is coupled to the output loadterminal 22.

Moreover, in this exemplary embodiment, one or more gallium nitridefield-effect transistors 32 feeds the matching network 28 an amplifiedcopy of a radio frequency signal input at the input terminal 20. Itshould be noted that a bias point for the peak amplifier 18 may be setfor other classes such as A or AB in order to configure the amplifierfor a desired response.

FIG. 2 is a schematic of a second embodiment of the load modulationamplifier 10 that is structured and configured in accordance with thepresent disclosure. Exemplary embodiments of the load modulationamplifier 10 may be fabricated using 0.15 micron T-gate gallium nitridehigh electron mobility transistor technology with a transition frequencygreater than 90 GHz. In the exemplary embodiment of FIG. 2, an amplifiedradio frequency signal that is output from the carrier amplifier 16 isinput into the third port P3 and undergoes a 90° phase shift beforeexiting the fourth port P4 that is coupled to the output load terminal22.

In some embodiments, the carrier amplifier 16 is biased with a firstsupply voltage and the peak amplifier 18 is biased with a second supplyvoltage, wherein the second supply voltage is between 10% and 50%greater than the first supply voltage. In some embodiments, the secondsupply voltage is between 50% and 100% greater than the first. In theexemplary embodiments of FIG. 1 and FIG. 2, the first supply voltage is10 V and the second supply voltage is 18 V, which is 80% greater thanthe first supply voltage. In yet other embodiments, the second supplyvoltage is between 100% and 200% greater than the first supply voltage.In yet other embodiments, the second supply voltage is between 200% and1000% greater than the first supply voltage.

FIG. 3 is an exemplary circuit layout for the matching network 28. Inthis example, the matching network is made of a first transmission lineTL1 and a second transmission line TL2 that are coupled in seriesbetween the field-effect transistors 32 and the network output terminal26. A first tuning stub ST1 and a second tuning stub ST2 are coupled toa node between the first transmission line TL1 and the secondtransmission line TL2. A first capacitor C1 is coupled between an outerend of the first tuning stub ST1 and a ground via pad 34-1. A dashedcircle represents a via. A second capacitor C2 is coupled between anouter end of the second tuning stub ST2 and a ground via pad 34-2. Inthis exemplary case, a tuning of the first tuning stub ST1 and tuning ofthe second tuning stub ST2 in combination with bias current levelsetting of the bias current IBIAS is carried out to ensure that outputimpedance Z0 of the peak amplifier 18 (FIGS. 1 and 2) at the networkoutput terminal 26 increases monotonically with increasing output powerof load modulation amplifier 10.

In this exemplary embodiment, the matching network 28 and thefield-effect transistors 32 are fabricated on a common substrate 36. Inthis example, the field-effect transistors 32 are individually labeledM1, M2, and M3. A radio frequency signal to be amplified is input at thegates G1, G2, and G3, respectively. Sources S1, S2, and S3 are coupledto ground via pads 34-3, 34-4, 34-5, and 34-6, respectively. Drains D1,D2, and D3 are coupled to the matching network 28 by way of a manifold38. It is to be understood that other combinations of matchingstructures and bias points that are realizable to accomplish themonotonically increasing peak amplifier output impedance Z0, so theexemplary embodiment of FIG. 3 is non-limiting.

An exemplary iterative design method employs radio frequency integratedcircuit (RFIC) simulation software that simulates performance of theembodiments of the load modulation amplifier 10. A goal of the exemplarydesign method is to ensure that output impedance Z0 of the peakamplifier 18 increases monotonically with increasing output power. Atleast another goal is to achieve efficient linear operation of the loadmodulation amplifier 10 at 10 dB output power backed off.

The exemplary iterative design method begins with choosing anappropriate circuit topology for the matching network 28. In anexemplary embodiment, an appropriate circuit topology is an L-networkmade up of microstrips. A model of the load modulation amplifier 10including the circuit topology of the matching network 28 is then inputinto a digital computer executing the RFIC simulation software. Nextsteps include setting initial values for components making up thecircuit topology and setting an initial current level for the biascurrent IBIAS. A simulation of the model of the load modulationamplifier 10 using the RFIC simulation software is invoked for thedesired output power range, which is swept to generate simulatedmeasurements of a scattering parameter S22 for the peak amplifier 18.Output impedance Z0 measurements may be derived from the scatteringparameter S22 measurements. Either the scattering parameter S22measurements or the output impedance Z0 measurements are processed todetermine if the output impedance Z0 increases monotonically withincreasing output power over a desired output power range that ensuresthe carrier amplifier is coupled at a lower output power level belowpredetermined power threshold value and decoupled at a higher outputpower level above the predetermined power threshold value.

If it is determined that the output impedance Z0 does not increasemonotonically with increasing output power over the desired output powerrange, at least one component value of the matching network is adjustedand/or the bias current IBIAS is adjusted before the output power rangeis sweep again to generate new simulated measurements of the scatteringparameter S22 for the peak amplifier 18. Either the new scatteringparameter S22 measurements or the new output impedance Z0 measurementsare processed to determine if the output impedance Z0 increasesmonotonically with increasing output power over the desired output powerrange. If the determination is positive, the method is complete and theload modulation amplifier 10 is realized and verified with laboratorytesting. Otherwise, the iterative design method continues until thesimulation indicates output impedance of the peak amplifier 18 increasesmonotonically with increasing output power at the output load terminal22. It is to be understood that the iterative design method of thepresent disclosure may be entirely automated by way of additionalprogram instructions executed by the digital computer that controlsexecution of the RFIC simulation software. Moreover, the programinstructions may adjust values for components making up the circuittopology and/or current level for the bias current IBIAS in a geneticfashion that converges to a desired level of output power linearity.

FIG. 4 is a graph of output impedance Z0 of peak amplifier 18 versusoutput power for the load modulation amplifier 10. The graph of FIG. 4shows an atypical yet highly desirable output impedance of the peakamplifier 18 that increases monotonically with increasing output powerat the output load terminal 22 over a wide output power range of between2.5 dBm and 35 dBm. In contrast, other types of amplifiers that have apeak amplifier such as a Doherty amplifier typically have decreasingoutput impedance with increasing power as the transistor of the peakamplifier dynamically biases up in current and do not have outputimpedance that increases monotonically with increasing output power.

In at least some exemplary embodiments of the load modulation amplifier10, the load modulation amplifier 10 provides linear voltage gain to aradio frequency signal having a frequency between 15 GHz and 100 GHz.Other exemplary embodiments of the load modulation amplifier 10 providelinear voltage gain to a radio frequency signal having a frequencybetween 30 GHz and 50 GHz.

The increase in output impedance of the peak amplifier 18 of the presentembodiments effectively steers power delivered to the output from thecarrier amplifier 16 to the peak amplifier 18. In at least one exemplaryembodiment, the output impedance of the peak amplifier 18 monotonicallyincreases between 30Ω and 100Ω with increasing output power between 2.5dBm and 35 dBm at the output load terminal. In at least one otherexemplary embodiment, the output impedance of the peak amplifiermonotonically increases between 30Ω and 50Ω with increasing output powerbetween 2.5 dBm and 29 dBm at the output load terminal. In at least oneadditional exemplary embodiment, the output impedance Z0 of the peakamplifier 18 monotonically increases between 50Ω and 100Ω withincreasing output power between 29 dBm and 35 dBm at the output loadterminal. In other embodiments, the output impedance Z0 of the peakamplifier 18 monotonically increases between 1.5 times and 4 times withincreasing output power over an output power back-off (OPBO) rangebetween 3 dB and 16 dB at the output load terminal

The graphs of FIGS. 5-8 are generated from simulations of the secondembodiment of the load modulation amplifier 10 of FIG. 2 with theisolation termination network 30 having isolation impedance that is lessthan 50Ω and specifically set at 0.1Ω for the simulations. FIG. 5 is agraph of carrier amplifier to output coupling versus millimeter wavefrequencies for the load modulation amplifier embodiments of the presentdisclosure. In particular, the graph of FIG. 5 shows that the carrieramplifier 16 decouples from the output load terminal 22 as outputimpedance Z0 of the peak amplifier 18 increases. Referring back to thegraph of FIG. 4, notice that at relatively low output power between 2.5dBm and 5 dBm the output impedance Z0 of the peak amplifier 18 is closeto 30Ω. As shown in FIG. 5 in solid line, the 30Ω output impedance Z0 ofthe peak amplifier 18 allows about a −3 dB coupling of the carrieramplifier's power to the output load terminal 22 over a frequency rangeextending from about 30 GHz to 50 GHz.

Referring back to the graph of FIG. 4, notice that at relativelymoderate output power close to 29 dBm the output impedance Z0 of thepeak amplifier 18 is close to 50Ω. As shown in FIG. 5 in dot-dash line,the 50Ω output impedance Z0 of the peak amplifier 18 allows only about a−4 dB coupling of the carrier amplifier's power to the output loadterminal 22 over a frequency range extending from about 36 GHz to 50GHz.

Referring back to the graph of FIG. 4 once again, notice that atrelatively high output power close to 32 dBm the output impedance Z0 ofthe peak amplifier 18 is close to 90Ω. As shown in FIG. 5 in dashedline, the 90Ω output impedance Z0 of the peak amplifier 18 allows onlyabout a −6 dB coupling of the carrier amplifier's power to the outputload terminal 22 over a frequency range extending from about 30 GHz to44 GHz. In the case of 90Ω output impedance Z0 of the peak amplifier 18,the contribution of the output power of the carrier amplifier 16 issmall enough that the carrier amplifier 16 may be considered practicallyde-coupled from the output load terminal 22. Such steering of outputpower from the carrier amplifier 16 to the peak amplifier 18 for arelatively low isolation impedance of 0.1Ω would not occur with aDoherty-type amplifier because Doherty operation would not provide anincrease in output impedance Z0 of the peak amplifier 18 as powerincreases.

FIG. 6 is a graph of carrier amplifier to output phase shift versusmillimeter wave frequencies for the load modulation amplifierembodiments of the present disclosure. Notice that carrier amplifieroutput phase shift through the output quadrature coupler 14 remainswithin ±20° of 0° between 30 GHz and 50 GHz for output impedance Z0 ofthe peak amplifier ranging between 30Ω and 90Ω.

FIG. 7 is a graph of peak amplifier to output coupling versus millimeterwave frequencies for the load modulation amplifier embodiments of thepresent disclosure. In particular, the graph of FIG. 7 shows that thepeak amplifier 18 couples to the output load terminal 22 as outputimpedance Z0 of the peak amplifier 18 increases. As shown in FIG. 7 insolid line, the 30Ω output impedance Z0 of the peak amplifier 18 allowsabout a −3 dB coupling of the peak amplifier's power to the output loadterminal 22 over a frequency range extending from 30 GHz to 50 GHz.Moreover, as shown in FIG. 7 in dot-dash line, the 50Ω output impedanceZ0 of the peak amplifier 18 yields a greater coupling of between justunder −2 dB and −2.5 dB of the carrier amplifier's power to the outputload terminal 22 over a frequency range extending from 30 GHz to 50 GHz.As further shown in FIG. 7 in dashed line, the 90Ω output impedance Z0of the peak amplifier 18 provides between about −2 dB and −2.5 couplingof the carrier amplifier's power to the output load terminal 22 over afrequency range extending from 30 GHz to 50 GHz. In the case of the 90Ωoutput impedance Z0 of the peak amplifier 18, the contribution of theoutput power at the output load terminal 22 of the load modulationamplifier 10 is large enough that the peak amplifier 18 may beconsidered practically coupled to the output load terminal 22.

FIG. 8 is a graph of peak amplifier to output phase shift versusmillimeter wave frequencies for the load modulation amplifierembodiments of the present disclosure. Notice that peak amplifier outputphase shift through the output quadrature coupler 14 remains within ±10°of 90° between 36 GHz and 42 GHz for output impedance Z0 of the peakamplifier ranging between 30Ω and 90Ω. In at least some embodiments, theload modulation amplifier 10 has a change in phase of no more than ±1°for output power over a 5 dB power range corresponding to half of agiven OPBO range.

FIG. 9 is a graph of power added efficiency and drain efficiency versusoutput power for the second embodiment of the load modulation amplifier10 of FIG. 2. The graph of FIG. 9 was generated by running a simulationof the load modulation amplifier 10 in which the carrier amplifier 16and the peak amplifier 18 were modeled with 0.15 micron T-gate galliumnitride high electron mobility transistor technology with a transitionfrequency greater than 90 GHz. A first supply voltage for the carrieramplifier was set to 10 V, while a second supply voltage for the peakamplifier was set to 18 V, and a first bias current of the carrieramplifier 16 was set to a level greater than a second bias current ofpeak amplifier 18 with the isolation termination network 30 being set to0.1Ω. These settings provided an OPBO drain efficiency of at least 45%at 10 dB output power back off. Power added efficiency depicted in thickdot-dash line shows an improvement of at least 6% at 10 dB OPBO over apower added efficiency depicted in thin dot-dash line for a conventionalDoherty amplifier modeled with the same 0.15 micron T-gate galliumnitride high electron mobility transistor technology with a transitionfrequency greater than 90 GHz. Moreover, drain efficiency depicted inthick solid line is improved by at least 8% at 10 dB OPBO over drainefficiency depicted in thin solid line for the conventional Dohertyamplifier modeled with the same 0.15 micron T-gate gallium nitride highelectron mobility transistor technology with a transition frequencygreater than 90 GHz.

FIG. 10 is a graph of third-order intermodulation distortion (IM3),third-order intercept point (IP3), and linearity figure of merit (LFOM)as a function of output power for the second embodiment of the loadmodulation amplifier 10 of FIG. 2. Note that the LFOM is equal to IP3divided by dissipated power. Solid lines represent responses of IM3,IP3, and LFOM for the first embodiment of the load modulation amplifier10 of FIG. 2, whereas dashed lines represent responses for IM3, IP3, andLFOM for a conventional Doherty amplifier modeled with the same 0.15micron T-gate gallium nitride high electron mobility transistortechnology with a transition frequency greater than 90 GHz. Notice thatthe load modulation amplifier 10 achieves greater than 10 dBc IM3performance over a wide output power range between 5 dBm and 23 dBm.Especially notice that the improvement in IM3 is much greater over theconventional Doherty at 22 dBm at 10 dB OPBO. Moreover, the loadmodulation amplifier 10 achieves an LFOM that is greater than 25:1 over10 dB OPBO, which is 5 times improved over the conventional Dohertyamplifier.

FIG. 11 is a graph of gain delta versus output power for amplitudemodulation-amplitude modulation (AM-AM) distortion for the secondembodiment of the load modulation amplifier 10 of FIG. 2 versus aconventional Doherty amplifier, and FIG. 12 is a graph of phase deltaversus output power for AM-phase modulation (AM-PM) distortion for thesecond embodiment of the load modulation amplifier 10 of FIG. 2 versus aconventional Doherty amplifier. As shown in the example of FIG. 11, thefirst embodiment has a change in amplitude gain of no more than 0.5% foroutput power between 15 dBm and 25 dBm over a 39-41 GHz frequency range.In at least some embodiments, the load modulation amplifier 10 has achange in amplitude gain of no more than 0.5% for output power over a 10dB power range corresponding to a given OPBO range. Both the loadmodulation amplifier 10 and the conventional Doherty amplifier weremodeled with the same 0.15 micron T-gate gallium nitride high electronmobility transistor technology with a transition frequency greater than90 GHz. Both the load modulation amplifier 10 and the conventionalDoherty amplifier simulations processed a complex wireless fidelity(Wi-Fi) signal similar to 802.11ac over 80 MHz of bandwidth with apeak-to-average power ratio between 9.0 dB and 9.5 dB. FIGS. 11 and 12give the resulting AM-AM and AM-PM distortion characteristics. Thickersolid lines represent 39 GHz, 40 GHz, and 41 GHz responses for the loadmodulation amplifier 10, whereas thinner solid lines represent 39 GHz,40 GHz, and 41 GHz responses for the conventional Doherty amplifier.Both of the graphs of FIG. 11 and FIG. 12 show that the load modulationamplifier 10 has relatively dramatic improvement over the conventionalDoherty amplifier.

For 5G millimeter wave systems, it is believed that an error vectormagnitude (EVM) lower than −26 dB may be adequate, which corresponds to˜5% EVM. FIG. 13 is a graph of EVM versus output power for the secondembodiment of the load modulation amplifier 10 of FIG. 2 in comparisonwith the conventional Doherty amplifier. Thicker solid lines represent39 GHz, 40 GHz, and 41 GHz responses for the load modulation amplifier10, whereas thinner solid lines represent 39 GHz, 40 GHz, and 41 GHzresponses for the conventional Doherty amplifier. Both the loadmodulation amplifier 10 and the conventional Doherty amplifier satisfythe linear requirement an EVM lower than −26 dB up to 22 dBm of outputpower for a 10 dB OPBO. However, the load modulation amplifier 10provides much lower distortion at higher OPBO levels, which illustratesan inherent linearity advantage over the conventional Doherty amplifier.For example, as shown in FIG. 13, the load modulation amplifier 10 hasan EVM of no more than 2% for an output power range between 10 dBm and20 dBm. In at least some embodiments, the load modulation amplifier 10has an error vector magnitude of no more than 2% for an output powercorresponding to a given OPBO range between 3 dB and 16 dB. Asmillimeter-wave communication systems evolve in the future, higher ordermodulation with higher peak to average power ratio will require lowerEVM requirements in the couple of percent or less to achieve improveddata throughput capability. As throughout, both the load modulationamplifier 10 and the conventional Doherty amplifier were modeled withthe same 0.15 micron T-gate gallium nitride high electron mobilitytransistor technology with a transition frequency greater than 90 GHz.

For millimeter wave communication systems, conventional power amplifiersare susceptible to a dynamically changing voltage standing wave ratio(VSWR), undesirable radio frequency interference, and variations inpackaging parasitic inductances and capacitances. As a result, the loadmodulation amplifier 10 is configured to counter such adverseconditions. FIG. 14 is a graph of a simulation result for EVM under a2:1 VSWR mismatch versus output power for the second embodiment of theload modulation amplifier 10 of FIG. 2. The graphed curves areassociated with effective antenna impedance due to antenna VSWRmismatch, which may be due to phased antenna array scanning and/or radiofrequency interference that may cause a lower than 50Ω VSWR mismatch ora higher than 50Ω VSWR mismatch. Vertical dashed lines indicate amaximum linear power range at which an EVM of −26 dB is met. The linearpower range from 5 dBm to 25 dBm over the 2:1 VSWR mismatch is arelatively drastic improvement over that illustrated in FIG. 15, whichis a graph of EVM under a 2:1 VSWR mismatch versus output power for aconventional Doherty amplifier. In at least some embodiments, the loadmodulation amplifier 10 has a maximum linear error vector magnitude foran output power corresponding to a given OPBO range between 3 dB and 16dB over a 2:1 voltage standing wave ratio mismatch. For the purpose ofcomparison, the conventional Doherty amplifier was modeled with the same0.15 micron T-gate gallium nitride high electron mobility transistortechnology with a transition frequency greater than 90 GHz that was usedto model the second embodiment of the load modulation amplifier 10 ofFIG. 2. Moreover, for both the second embodiment of the load modulationamplifier 10 and the conventional Doherty amplifier, a first supplyvoltage for the carrier amplifier 16 was set to 10 V, while a secondsupply voltage for the peak amplifier 18 was set to 18 V, and a firstbias current the carrier amplifier 16 was set to a current level greaterthan a second bias current of the peak amplifier 18 with the isolationtermination network 30 being set to 0.1Ω.

The embodiments of the present disclosure are employable in fundamentallinear efficient gallium nitride power amplifier applications. Suchapplications include, but are not limited to, 5G base stations, 5Gmillimeter phased arrays, Wi-Fi 802.11ax, CATV DOCSIS 3.1 Plus, andadvanced military and defense radio communications.

Those skilled in the art will recognize improvements and modificationsto the preferred embodiments of the present disclosure. All suchimprovements and modifications are considered within the scope of theconcepts disclosed herein and the claims that follow.

1. A load modulation amplifier comprising: a carrier amplifier foramplifying a radio frequency signal when input power of the radiofrequency signal is below a predetermined power threshold value; a peakamplifier coupled in parallel with the carrier amplifier for amplifyingthe radio frequency signal when input power of the radio frequencysignal is above the predetermined power threshold value, wherein thecarrier amplifier is biased with a first supply voltage and the peakamplifier is biased with a second supply voltage that is between 10% and1000% greater than the first supply voltage; and an output quadraturecoupler configured to combine power from both the carrier amplifier andthe peak amplifier for output through an output load terminal, whereinoutput impedance of the peak amplifier monotonically increases withincreasing output power at the output load terminal.
 2. The loadmodulation amplifier of claim 1 wherein output impedance of the peakamplifier monotonically increases between 30Ω and 100Ω with increasingoutput power between 2.5 dBm and 35 dBm at the output load terminal. 3.The load modulation amplifier of claim 1 wherein output impedance of thepeak amplifier monotonically increases between 30Ω and 50Ω withincreasing output power between 2.5 dBm and 29 dBm at the output loadterminal.
 4. The load modulation amplifier of claim 1 wherein outputimpedance of the peak amplifier monotonically increases between 1.5times and 4 times with increasing output power over an output powerback-off range between 3 dB and 16 dB at the output load terminal. 5.The load modulation amplifier of claim 1 wherein the output quadraturecoupler is a Lange coupler.
 6. The load modulation amplifier of claim 1wherein the peak amplifier is configured to couple to an isolationtermination network having impedance greater than 50Ω and the carrieramplifier is coupled to the output load terminal through 0° phase shiftports of the output quadrature coupler.
 7. The load modulation amplifierof claim 1 wherein the peak amplifier is configured to couple to anisolation termination network having impedance less than 50Ω, whereinthe carrier amplifier is coupled to the output load terminal through 90°phase shift ports of the output quadrature coupler.
 8. The loadmodulation amplifier of claim 1 wherein the carrier amplifier is biasedwith the first supply voltage and the peak amplifier is biased with thesecond supply voltage, wherein the second supply voltage is between 10%and 50% greater than the first supply voltage.
 9. The load modulationamplifier of claim 1 wherein the carrier amplifier is biased with thefirst supply voltage and the peak amplifier is biased with the secondsupply voltage, wherein the second supply voltage is between 50% and100% greater than the first supply voltage.
 10. The load modulationamplifier of claim 1 wherein the carrier amplifier is biased with thefirst supply voltage and the peak amplifier is biased with the secondsupply voltage, wherein the second supply voltage is between 100% and1000% greater than the first supply voltage.
 11. The load modulationamplifier of claim 8 wherein bias current provided to the peak amplifieris less than bias current provided to the carrier amplifier.
 12. Theload modulation amplifier of claim 1 wherein the carrier amplifier isbiased with the first supply voltage and the peak amplifier is biasedwith the second supply voltage that is different from the first supplyvoltage, wherein bias current provided to the peak amplifier is lessthan bias current provided to the carrier amplifier.
 13. The loadmodulation amplifier of claim 1 wherein the peak amplifier comprises amatching network that provides the output impedance of the peakamplifier that increases with increasing output power at the output loadterminal.
 14. The load modulation amplifier of claim 13 wherein thematching network comprises only passive electrical components.
 15. Theload modulation amplifier of claim 1 wherein the peak amplifier is agallium nitride amplifier.
 16. The load modulation amplifier of claim 1wherein the load modulation amplifier provides linear voltage gainbetween 15 GHz and 100 GHz.
 17. The load modulation amplifier of claim 1wherein the load modulation amplifier provides linear voltage gainbetween 30 GHz and 50 GHz.
 18. The load modulation amplifier of claim 1having a change in amplitude gain of no more than 0.5% for output powerover a 10 dB power range corresponding to a given output power back-offrange.
 19. The load modulation amplifier of claim 1 having a change inphase of no more than ±1° for output power over a 5 dB power rangecorresponding to half of a given output power back-off range.
 20. Theload modulation amplifier of claim 1 having an error vector magnitude ofno more than 2% for an output power corresponding to a given outputpower back-off range between 3 dB and 16 dB.
 21. The load modulationamplifier of claim 1 having maximum linear error vector magnitude for anoutput power corresponding to a given output power back-off rangebetween 3 dB and 16 dB over a 2:1 voltage standing wave ratio mismatch.